This is the accepted version of the article: Herrojo, Cristian; Paredes, Ferran; Mata Contreras, Francisco Javier. «Double- stub loaded microstrip line reader for very high data density microwave encoders». IEEE Transactions on microwave theory and techniques, Vol. 67 , issue 9 (Sept. 2019), p. 3527-3536. DOI 10.1109/TMTT.2019.2929128 This version is avaible at https://ddd.uab.cat/record/221405 under the terms of the license Double-Stub Loaded Microstrip Line Reader for Very High Data Density Microwave Encoders Cristian Herrojo, Member IEEE, Ferran Paredes, Member IEEE, and Ferran Martín, Fellow IEEE Abstract— Compact and high-data density microwave Microwave encoders are of interest for motion control encoders useful for motion control and near-field chipless- applications, and for the implementation of chipless RFID applications are proposed in this paper. The encoders radiofrequency identification (chipless-RFID) systems. In are chains of metallic strips etched on a dielectric substrate. the former application, the relative velocity and The reader consists of a microstrip line loaded with a pair of identical open-ended folded stubs located at different positions displacement between the encoder and the reader is and oriented face-to-face by their extremes. By displacing the determined from the time distance between adjacent peaks, encoder over the extremes of the stubs, inter-stub coupling or dips, provided the chain period is known and all the arises when a strip is located on top of the stubs, thereby metallic elements are present (and are functional) at the generating two transmission zeros (rather than one) in the predefined chain positions. Both angular and linear frequency response of the line. Thus, the presence of a strip on displacement and velocity sensors based on microwave top of the face-to-face stubs produces a variation in the transmission coefficient of the line, which in turn can be encoders (either circularly or linearly shaped) have been detected by feeding the line with a harmonic signal, recently reported [4]-[8]. conveniently tuned. Encoder motion generates an amplitude For identification purposes, certain metallic elements of modulated (AM) signal at the output port of the line with the encoder chain are either eliminated or made inoperative. peaks, or dips, separated a time distance dictated by the By this means, an ID code is generated, and the encoder can relative velocity between the reader and the encoder. be used as a chipless-RFID tag, where the encoded chain Moreover, by making certain strips of the chain inoperative replaces the application specific integrated circuit (ASIC), (e.g., by cutting them), it is possible to encode information that can be read as the absence (logic state ‘1’) or presence (logic or chip, present in chipped-RFID tags. In these chipless- state ’0’) of peaks, or dips, at predefined positions in the RFID systems, tags are read by proximity (through near output (AM) signal of the reader line. Since short strips suffice field) by displacing them over the reader, following a time- to generate inter-stub coupling, unprecedented data density division multiplexing scheme where the bits are read per surface (DPS = 26.04 bit/cm2) is obtained, as revealed by sequentially [8]-[17]. The advantage of this unconventional the implementation of 6.4 mm x 60 mm 100-bit encoders. time-domain chipless-RFID approach is the data storage Index Terms– Chipless-RFID, microstrip technology, microwave encoder, motion control. capacity, only limited by the encoder (or tag) size. Thus, chipless-RFID tags with unprecedented number of bits have I. INTRODUCTION been reported [8],[14],[16]. This number of bits, comparable to the one of chipped tags, is by far superior to M ICROWAVE ENCODERS can be implemented by etching or printing metallic elements (e.g., resonators, or linear strips) at predefined positions in a dielectric substrate, the number of bits reported in the available literature relative to full planar chipless-RFID tags (either based on time-domain [18]-[31], frequency-domain [32]-[52], or typically arranged forming linear or circular chains. The exploiting various domains simultaneously [53]-[64]). working principle of such encoders is based on the A figure of merit in microwave encoders for its use as detection of the presence or absence of metallic elements in linear or angular displacement and velocity sensors is the the chain (revealed as peaks or dips in the output signal) chain period, intimately related to encoder resolution. In when the encoder is displaced over a dedicated reader. [8],[16] microwave encoders with a period of 0.6 mm were Indeed, this working principle is similar to the one of reported. The encoder period is related to the information optical encoders [1]-[3], implemented as chains of apertures density per unit length (DPL), a key parameter when the in a metallic screen. However, whereas in optical encoders encoder is used as near-field chipless-RFID tag. Note that the presence/absence of apertures at predefined positions is such period corresponds to a per-unit-length data density of detected by means of an optical beam, in microwave DPL = 16.7 bit/cm [8],[16], i.e., superior to the one encoders the presence or absence of metallic elements is achieved in [15] (DPL = 5 bit/cm), where the chain period sensed by means of an electromagnetic signal, e.g., through of the encoders was 2 mm. near-field coupling. However, in chipless-RFID tags, the data density per surface (DPS) is also important, since it determines the area of the tag (once the number of bits is set to a certain value) This paper is an expanded version from the IEEE Latin America Microwave Conference 2018, Arequipa, Peru, December 12-14, 2018. and consequently tag cost. In [8], [15] and [16], the This work was supported by MINECO-Spain (project TEC2016-75650- achieved data density per surface is DPS = 7.44 bit/cm2, R), by Generalitat de Catalunya (project 2017SGR-1159), by Institució DPS = 1.67 bit/cm2, and DPS = 4.9 bit/cm2, respectively. Catalana de Recerca i Estudis Avançats (who awarded Ferran Martín), and These values are good, but the microwave encoders by FEDER funds. C Herrojo, F. Paredes and F. Martín are with GEMMA/CIMITEC, presented in [8],[15],[16] use metallic elements consisting Departament d’Enginyeria Electrònica, Universitat Autònoma de of straight strips with excessive length, dictated by the Barcelona, 08193 Bellaterra, Spain. E-mail: Ferran.Martin@uab.es. frequency of the interrogation signal. In [15],[16], the Reader Circulator Envelope encoder strips act as half-wavelength resonators, whereas detector Δx/vr/Tag ID Z0 … V0 (t) V0 the reader is a single stub-loaded line. In [8], the encoder ... Vi @ f0 Z0 R C strips do not behave as resonant element. However, in order t0 t0 tN-1 to detect strip functionality (or presence/absence), the … tN-1 reader uses a half-wavelength resonator, and the length of the strips necessary for their detectability should be Encoder comparable to the dimensions of such reader resonator. Fig. 1. Sketch of the proposed reader-encoder system. Increasing the frequency of the interrogation signal provides a way to reduce the length of the strips, but at the expense of increasing the complexity of the reader a chain of metallic elements (the encoder) when such chain electronics. is displaced on top of the reader line at short distance. In this paper, we propose a new encoder-reader system, Encoder motion modulates the transmission coefficient of inspired by the work first presented in [15] and then the line (through electromagnetic coupling), resulting in an optimized in [16], where the reader is a microstrip line amplitude modulated (AM) signal at the output port of the loaded with a pair of open-ended folded stubs (oriented line, when it is fed by a harmonic (single tone) signal face-to-face by their extremes), and the encoders are chains conveniently tuned. With this system, the information in of very short straight strips oriented orthogonally to the regard to the relative displacement or velocity between the chain axis. The encoder strips act as narrow patch reader and the encoder is contained in the envelope capacitors able to enhance the coupling between the stubs function, with peaks, or dips, separated a time distance when such strips are located on top of the open end of the dictated by such velocity. For near-field chipless-RFID stubs. The effect is a variation in the transmission applications, where some elements of the encoder (i.e., coefficient of the double stub loaded line, which is used to those corresponding to the logic state ‘0’) are made detect the presence/absence of functional strip in the chain inoperative, the ID code is also contained in the envelope (as it will be later discussed). function. In this case, the relative velocity between the The working principle of the presented encoder-reader reader line and the encoder must be well known (for that system is similar to the one reported in [8],[15],[16]. purpose, header bits may be used). Tag reading is However, the encoder strips of this new system can be performed at predefined time windows, with either maxima much shorter, provided they merely act as capacitive or minima in the envelope function dictated by the presence elements (rather than resonant elements), able to modulate (logic ‘1’) or absence (logic ‘0’) of functional metallic inter-stub coupling. Moreover, with the proposed reader, element on top of the reader line. To obtain the envelope exhibiting a frequency response with a movable pole and function, an envelope detector is needed. The output data transmission zero (or zeros), a large excursion of the can be visualized in an oscilloscope [11] or, alternatively, transmission coefficient at the frequency of the such data can be post-processed (using a data acquisition interrogation signal is achievable, as it will be shown. This card) [14] in order to infer either the ID code or the is a fundamental aspect, with direct impact on system displacement/velocity. robustness against misalignment (between the encoder and In this system, encoder reading proceeds by feeding the the reader) and vibrations. Thus, the proposed system reader line with a harmonic interrogation signal, contrary to represents a significant progress as compared to previous most chipless-RFID systems, where wideband signals are encoder-reader systems based on encoders implemented by typically required [18]-[64]. This aspect is important since means of straight strips [8],[15],[16]. the cost of the associated electronics of the reader can be The work is organized as follows: the working principle reduced. Namely, neither narrow pulses (as needed in time- of the proposed system is explained in detail in Section II, domain reflectometry based chipless-RFID systems where system needs are also pointed out. Section III is [21],[22]), nor sweeping interrogation signals covering a devoted to analyze the proposed reader-encoder system, in wide spectral bandwidth (as used to read spectral signature order to justify the chosen reader topology, based on a barcodes [32]-[35]), are involved in the proposed system. double stub loaded line, in terms of encoder size, data The frequency of the interrogation signal, f0, must be capacity and density, and system robustness. In Section IV, chosen so that a large excursion of the transmission the system is experimentally validated, whereas a coefficient at that frequency (when the logic state switches) comparison to other approaches is carried out in Section V. occurs. A significant variation of the transmission A tolerance analysis is carried out in Section VI. Finally, coefficient between the upper and lower value is necessary the main conclusions are highlighted in Section VII. in order to obtain a high modulation index, which in turn provides system robustness against misalignments and II. WORKING PRINCIPLE AND SYSTEM NEEDS vertical distance (air gap) variations between the reader and the encoder. However, it is also necessary that the The working principle of the proposed system is depicted magnitude of the upper value of the transmission coefficient in Fig. 1 (it is similar to the one of other near-field is close to 0 dB, in order to guarantee a minimum chiplessRFID systems [8]-[17] and displacement/velocity (detectable) level of the envelope function. Finally, it is sensors [4]-[7], but it is reproduced here for coherence and convenient that the transmission coefficient of the reader completeness). The essential (sensitive) part of the reader is line exhibits as much vertical transitions as possible, with a transmission line based element sensitive to the effects of large frequency variation when the logic states switches (a) 0 (b) 0 l3 f0 strips |S21| (dB) |S21| (dB) P Pin |S21| |S21| -10 -10 l1 l4 W -20 -20 fL fH fL fH … ls s Reader l2 frequency window frequency window (c) 0 l5 (d) -10 |S21| ws Pout |S21| (dB) |S21| (dB) ws |S21| -5 -20 Encoder |S21| unloaded Fig. 3. Topology of the proposed reader-encoder and relevant dimensions -10 |S21| loaded -30 (in mm). W = 1.81, l0 = l2 + ws = 2.2, l1 = 3.8, l3 = 3.3, l4 = 2.1, l5 = 2.1, fL fH fL fH s = 1.60, P = 0.60, ws = 0.20 and lS = 6.4. frequency window frequency window 0 EM sim. bare reader Circuit sim. bare reader (C = 0.5 fF) 200 S11 Fig. 2. Illustration of various situations of frequency responses of reader- -20 encoder for both logic states (‘0’ and ‘1’). (a) Large transmission S21 0 21 coefficient excursion and wide frequency window with high transmission -40 coefficient for the upper level; (b) large transmission coefficient excursion fp f 11 -200 but narrow frequency window with high transmission coefficient for the -60 z upper level; (c) small transmission coefficient excursion and wide 0 |S11|, |S21| (dB) EM sim. reader loaded (air gap = 0.2 mm) 200 11, 21 ( ) Circuit sim. reader loaded (C = 5.5 fF) frequency window with high transmission coefficient for the upper level; S11 (d) large transmission coefficient excursion and wide frequency window -20 with low transmission coefficient for the upper level. S21 21 0 -40 between ‘0’ and ‘1’, or vice versa. This aspect results in -60 11 -200 better system robustness against fabrication related 0 EM sim. reader loaded (air gap = 0.15 mm) tolerances of both the line and the reader. Circuit sim. reader loaded (C = 8 fF) 200 S11 To further illustrate system needs, Fig. 2 schematically -20 S21 21 0 depicts various scenarios corresponding to different -40 frequency responses of hypothetical readers-encoders for 11 -200 both logic states (i.e., with the sensitive part of the reader -60 3.6 4.0 4.4 3.6 4.0 4.4 loaded with either functional or inoperative metallic Frequency (GHz) element of the encoder). According to the previous words, Fig. 4. Simulated frequency response (without losses) of the bare reader the preferred case is the one of Fig. 2(a). and reader loaded with a strip in the sensitive region. Different air gaps are considered, corresponding to different coupling capacitances (indicated). III. THE PROPOSED READER-ENCODER SYSTEM AND The electromagnetic simulations have been carried out with Keysight Momentum. ANALYSIS A reader-encoder structure exhibiting a frequency transmission zeros is visible, and the pole is displaced to response of the type shown in Fig. 2(a), compatible with lower frequencies. The splitting in the transmission zeros is high data density encoders is depicted in Fig. 3. The reader, indicative of certain level of coupling between the stubs, in black color, is a 50- microstrip line loaded with a pair which is not totally negligible for the bare reader. of open-ended folded stubs separated a distance l0. The Nevertheless, it is clearly manifested when a strip is located encoder, in grey color, is a chain of transversally oriented on top of the stubs, by virtue of the significant separation short strips etched on a dielectric substrate. When an between the transmission zeros (note that for uncoupled encoder strip is located on top of the open-ended extremes stubs, the single transmission zero corresponds to the of the stubs (the sensitive region), capacitive coupling frequency where the length of the stubs is a quarter between the stubs arises, resulting in a modification of the wavelength, or odd multiple). transmission coefficient useful for our purposes. Interestingly, the pole position in the frequency response Figure 4 depicts the responses corresponding to a bare of the bare reader roughly coincides with the first reader (i.e. without encoder strip in the sensitive region) and transmission zero of the response of the loaded reader (for a reader loaded with a strip. The considered substrate for the vertical distance between the reader and the encoder, or air reader is the Rogers RO4003C with thickness h = 0.81 mm, gap, of 0.2 mm). By this means, a large excursion in the dielectric constant r = 3.55, and dissipation factor transmission coefficient at the pole, or transmission zero, tan = 0.0021. For the encoder, the considered substrate is frequency results. Thus, the frequency of the interrogation the same as the reader but with thickness h = 0.203 mm. It signal, f0, must be set to a value in the vicinity of those can be seen that for the bare reader, the response exhibits a frequencies. Obviously, the variation experienced by the pole and two closed transmission zeros. By contrast, for the transmission coefficient when a strip is located in the strip-loaded reader, further frequency splitting in the sensitive region of the reader depends on the air gap Z0, Φ0 with (a) Pin Pout 0 𝑌𝑖𝑛𝑒 = 𝑗 [𝑌0 tan + 𝑌1 tan 1 ] (2a) 2 𝑌 1+21𝐶 tan 1 Z1, Φ1 Z1, Φ1 symmetry 0 plane 𝑌𝑖𝑛𝑜 = −𝑗 [𝑌0 cot + 1 ] (2b) 2 𝑍1 tan 1 − 2𝐶 In (1) and (2),  is the angular frequency, Y0 = 1/Z0 and 2C 2C Y1 = 1/Z1 are the line admittances, and Yine and Yino are the admittances seen from the input ports in the even- and odd- (b) Even-mode (c) Odd-mode mode circuits, respectively. The transmission and reflection coefficients of the structure of Fig. 5(a) are simply given by Z0, Φ0/2 Z0, Φ0/2 Pin Pin [65] 1 𝑆21 = 𝑆12 = (𝑒 − 𝑜 ) (3a) 2 Z1, Φ1 Z1, Φ1 1 𝑆11 = 𝑆22 = (𝑒 + 𝑜 ) (3b) 2 Let us now consider that C = 0 pF, corresponding to 2C uncoupled stubs. In this case, the transmission coefficient Fig. 5. Circuit schematic of the proposed reader-encoder system (a), and exhibits a single transmission zero located at the frequency equivalent circuit for even-mode (b) and odd-mode (c) excitation. where 1 = /2 (and at the odd multiples). This is straightforward by inspection of the circuit model of separation. By further reducing such distance, inter-stub Fig. 5(a) with C = 0 pF. Nevertheless, this conclusion is coupling increases, resulting in a progressively enhanced also inferred analytically, since for 1 = /2 it follows that frequency window in the responses, as it can be seen in Fig. e = o = −1, and consequently S21 = S12 = 0. As expected, 4. In practice, it is difficult to accurately control the air gap the electrical length of the transmission line section between separation. For that reason, it is convenient to design the the stubs, 0, does not have any influence on the reader-encoder with an appreciable frequency window for transmission zero position when C = 0 pF. However, 0 the nominal value of the air gap distance. determines the location of the pole. Such pole position The positions of the pole, p = 2fp, and transmission depends also on 1, but this parameter is dictated by the zero(s), z = 2fz, determined by the distance between stubs, position of the transmission zero. In other words, the length l0, stub length, l1, and coupling level, are important for of the stubs is not considered to be a design parameter, design purposes. To gain insight on this, let us calculate the since it is fixed by the transmission zero of the bare transmission and reflection coefficient of the proposed structure, a system specification. The pole arises at that structure, modeled by the schematic depicted in Fig. 5. In frequency satisfying S11 = S22 = 0, or e + o = 0. Since the that figure, C models the coupling capacitance between the circuits of Figs. 5(b) and (c) are purely reactive, i.e., e = stubs, 0 and 1 are the electrical length of the transmission o = 1, it follows that e and o should exhibit angles line section (between the stub positions) and the electrical differing in  (or an odd multiple) at the frequency of the length of the stubs, respectively, and Z0 and Z1 are the pole. According to (1) and (2), with C = 0 pF, this condition corresponding characteristic impedances (with Z0 = 50 , can be expressed as: as mentioned before). The model is validated by simulating the frequency response with the circuit simulator of 0 𝑌1 Keysight ADS. The single parameter that has been adjusted 2 arctan (− tan − tan 1 ) − 2 𝑌0 in order to fit the electromagnetic simulations of Fig. 4 is the coupling capacitance. It can be seen (Fig. 4) that the 0 𝑌1 agreement between the electromagnetic and circuit −2 arctan (cot − tan 1 ) = (2𝑛 + 1) (4) 2 𝑌0 simulations is excellent. Thus, it is demonstrated that the effect of strips on top of the sensitive region of the reader where n is an integer. can be modeled through capacitive coupling between the In (4), the ratio Y1/Y0 is not considered to be a design stubs. parameter. The reason is that Z0 = 1/Y0 = 50 , as indicated Since the reader line is symmetric with regard to the bi- before, and Y1 is determined by the width of the stubs. Such section plane between the ports, the S-parameters of the width is set to the same value than the width of the encoder structure can be inferred from the well-known even-odd strips. By this means, inter-stub coupling caused by mode analysis [65]. The equivalent circuits for even- and multiple strips simultaneously is reduced. On the other odd-mode excitations are also depicted in Fig. 5. The hand, the electrical length of the stubs can be expressed as reflection coefficients for the even- and odd-mode 𝑙 equivalent circuits are 1 = 𝑙1 0 (5) 𝑒 0 𝑌0 −𝑌𝑖𝑛 𝑒 = 𝑌 +𝑌𝑒 (1a) where it has been considered that the phase velocities (or 0 𝑖𝑛 the phase constants) of the line and stubs are identical (a 𝑌 −𝑌 𝑜 𝑜 = 𝑌0+𝑌𝑖𝑛𝑜 (1b) reasonable approximation). Thus, expression (4) depends 0 𝑖𝑛 on a single variable, 0, provided the ratio l1/l0 is set to a certain value. In other words, 0 (and consequently the frequency of the pole) is dictated by the ratio l1/l0. However, 1.6 1.6 it is not possible to isolate 0 from (4). Thus, we have obtained numerically 0 as a function of l1/l0 (see Fig. 6). fi0new 1.2 1.2 From the value of 0, the ratio between the pole and p/z = 0.962 0 (rad)  p/  z transmission zero frequencies is given by 0.8 0.8 𝑝 20 𝑙1 = (6) 𝑧  𝑙0 0.4 0.4 0 = 0.312 rad and this ratio is also depicted in Fig. 6. Thus, the relative 0.0 0.0 position between the pole and the zero for the case of 0 2 4 6 8 10 uncoupled stubs (C = 0 pF), is determined by the ratio l1/l0 between the stub length and transmission line section Fig. 6. Dependence of 0 and p/z on l1/l0. length. With the pole for the bare reader (C = 0 pF) set to a (a) certain value, p, it is convenient to locate the first transmission zero frequency, z’, for the strip-loaded reader (C  0 pF) at p (in order to obtain a significant excursion in the transmission coefficient, as discussed before). According to (1) and (2), the condition for the transmission Strips for guiding stubs the encoder zero frequency with C  0 pF is 0 𝑌1 2 arctan (− tan − tan 1 ) − 2 𝑌0 𝑌 1 0 1 1+2𝐶 tan 1 −2 arctan (cot + ) =0 (7) 2 𝑌0 𝑍1 tan 1 − 1 (b) 2𝐶 and it depends on C. Thus, C must be set to the value providing z’ = p. Since at this frequency ( z’ or p), 0 and 1 are known, the single unknown in expression (7) is C, and this parameter can be obtained numerically. The design of the reader-encoder depicted in Fig. 3 has been aided with the previous procedure. First, the transmission zero frequency of the bare structure and by considering completely uncoupled stubs (C = 0 pF) has been set to fz = z/2 = 4.03 GHz. This has provided the Fig. 7. Photograph of the fabricated reader (a) and 100-bit encoder with all bits set to ‘1’ logic state (b). length of the stubs (l1 = 11.3 mm in the considered substrate). The stub width has been set to ws = 0.2 mm, strips, the data capacity per surface area achieved in this identical to the width of the strips, for the reasons work is DPS = 26.04 bit/cm2, i.e., 3.50, 15.6 and 5.31 times mentioned before. With the considered reader substrate, the larger than those reported in [8], [15] and [16], respectively. characteristic impedance of the stub lines has been found to be Z1 = 133 . Then, the position of the pole for the bare IV. EXPERIMENTAL VALIDATION reader has been set to fp = p/2 = 3.88 GHz, resulting in p/ z = 0.962. With this ratio, the inter-stub distance, l0, has The photograph of the designed and fabricated reader and been inferred from the results depicted in Fig. 6, providing the one of a 100-bit encoder with all the strips present, and l0 = 2.33 mm (and hence l1/l0 = 4.85). Finally, using (7), the functional, at the predefined positions are shown in Fig. 7. capacitance C has been calculated in order to obtain z’ = The reader and the tags have been fabricated by means of p. The resulting value, C = 4.5 fF, coincides to a good the milling machine LPKF H100. The measured response of approximation with the value of C given in Fig. 4, the bare reader, inferred with the Agilent N5221A vector corresponding to the air gap case providing z’ = p. network analyzer, is depicted in Fig. 8, where it is It is important to highlight that with the nominal air gap compared with the electromagnetic simulation including value, the encoder strip length necessary to achieve the losses. Such figure also includes the response of the reader required capacitance C is as small as 0.2 mm. Taking into with a strip on top of it. The agreement between the lossy account that the strip width and inter strip distance is electromagnetic simulation and the measured responses is 0.2 mm and 0.4 mm, respectively, the chain period is 0.6 very reasonable, taking into account the difficulty in mm. With these geometrical variables of the encoder, the controlling the air gap distance (nominally set to 0.2 mm) information density per unit length is DPL = 16.7 bit/cm, with our experimental set up. identical to the one reported in [8],[16], and much better The photograph of the experimental setup used to obtain than the one reported in [15] (with DPL = 5 bit/cm). the envelope function of the different considered encoders However, due to the substantially reduced length of the in response to the interrogation signal is depicted in Fig. 9. 0 f0 Isolator Schottky |S21| (dB) diode -20 V0 -40 Envelope detector 0 Reader Encoder |S11| (dB) -20 EM sim. bare reader Meas. bare reader EM sim. reader loaded with an encoder Meas. reader loaded with an encoder Vi @ f0 -40 Fig. 9. Photograph of the experimental setup. 3.6 3.8 4.0 4.2 4.4 Frequency (GHz) 2.0 (a) 1.0 V0/Vmax Fig. 8. Measured and simulated responses of the bare reader and reader 0.5 1.5 loaded with an encoder strip on top of it. 0.0 1 1 0 1 1 0 Such setup includes a function generator (providing the 1.0 interrogation signal), an envelope detector preceded by an isolator (to avoid mismatching reflections from the diode), 0.5 and an oscilloscope, where the envelope functions of the different tags are visualized. Such setup is identical to the 0.0 one described in [11], where details of the specific equipment and electronic components are provided. The 2.0 (b) 1.0 V0/Vmax single difference concerns the sensitive part of the reader, a 0.5 double-stub loaded line (see Fig. 7) connected between the 1.5 V0/Vmax output port of the function generator and the input port of 0.0 1 0 1 0 1 0 the detector. A guiding system is used to displace at 1.0 controllable distance the encoders over the sensitive part of the reader. The accuracy and robustness of such guiding 0.5 system against air gap variation is limited. Nevertheless, the different encoders can be correctly read, as it will be shown 0.0 next. Three different 100-bit encoders have been used to 2.0 (c) 1.0 vm = 5P/Tm = 1.46 cm/s V0/Vmax validate the proposed approach. In all the cases, a periodic 0.5 Tm = 0.205 s ID code has been considered, particularly, one with all bits 1.5 set to ‘1’ logic state, i.e., ‘11111…’, one with the sequence 0.0 1 1 1 1 1 1 ‘101010…’, and one with the sequence ‘110110110…’. 1.0 Rather than fabricating three different encoders, we have opted to program two of them after being fabricated with all 0.5 bits set to ‘1’ logic state. Namely, we have cut those strips corresponding to the logic state ’0’. By this means, these 0.0 strips are made inoperative, and roughly the same effect as 1.5 3.0 4.5 the absence of strip on top of the sensitive part of the reader Time (s) is achieved. The envelope functions of these encoders are depicted in Fig. 10. Measured normalized envelope function corresponding to the indicated codes. Fig. 10, where it can be appreciated that the different ID codes are correctly read. The interrogation signal frequency However, it can be appreciated that the minimum of the has been set to f0 = 3.88 GHz, the frequency of the pole for envelope function does not appear exactly at the time when the bare reader. The dynamic margin, or separation between perfect overlapping between the encoder strip and the the voltage levels corresponding to the two states, is high. reader occurs. This is explained by an under air gap distance. According to Fig. 4, if the gap is below the nominal value, the coupling capacitance increases, and the encoders occupy an area as small as 0.64 cm  6 cm. This response is over-shifted to lower frequencies. The result is surface density, a figure of merit in chipless-RFID tags, is that the first transmission zero of the strip-loaded reader no 3.50, 15.6 and 5.31 times larger than those reported in [8], longer coincides with the pole of the bare reader (where the [15] and [16], respectively, where encoders based on strip interrogation signal is tuned). This reduces the dynamic chains are also considered. As compared to other encoders margin, but there is a relative position between the reader based on chains of resonant elements [4]-[7],[9]-[14], the and strip where the first transmission zero coincides with surface density obtained in the present work is significantly the pole of the bare reader (or interrogation signal superior (e.g., the best DPS achieved in resonator-based frequency). At this position, the envelope function reaches encoder is the one reported in [11], with a value of its minimum value. In view of Fig. 10 (where clearly DPS = 8.4 bit/cm2). separated voltage levels between the binary states can be If the comparison is extended to other chipless-RFID appreciated), the functionality of the proposed approach for systems based on full planar encoders readable in time or the implementation of near-field chipless-RFID systems is frequency domain [18]-[64], to the best of our knowledge, validated. the reported DPS and data capacity (number of bits) in The functionality of the system for the measurement of those works are far from the values achieved in this work linear velocities and displacements is apparent in view of (as an example, a tag with DPS = 5.88 bit/cm2 and data Fig. 10(c). The time period of the envelope function for the capacity of 64 bits is reported in [55]). Note that the number ‘11111…’ code (easily detectable, e.g., by means of a post- of bits (100 in the fabricated encoders) is only limited by processing system, out of the scope of this paper) is given tag size, as far as tag reading proceeds by time-division by the time lapse between two crosses of adjacent strips multiplexing using a harmonic interrogation signal. Further above the sensitive part of the reader. Since the spatial increasing the number of bits is compatible with acceptable period is well known (0.6 mm in our case), the relative tag dimensions and does not represent an additional instantaneous velocity between the encoder and the reader complexity in the reader. In summary, an unprecedented can be easily inferred. For the envelope function of combination of number of bits and data density has been Fig. 10(a), the velocity is found to be 1.46 cm/s (actually, achieved with the proposed system. the instantaneous velocity slightly varies, so that such speed It is also remarkable the fact that the interrogation signal value has been obtained by averaging over 5 periods). The in the proposed system is a harmonic (single-tone) signal, relative displacement between the reader and the encoder rather than a wideband pulsed signal, or a sweeping from a reference position is simply given by the cumulative frequency signal, as required in most time domain and number of peaks, or dips, in the envelope function from that frequency-domain, respectively, chipless-RFID systems position. [18]-[64]. This has direct impact in the cost associated to the reader electronics, which is reduced if wideband signals V. COMPARISON TO OTHER APPROACHES are not involved. However, the reported system requires The proposed system features a space resolution, which is encoder reading by proximity and proper alignment with the dictated by the encoder period, i.e., 0.6 mm. The space reader. Whereas in motion control applications this cannot resolution in the system reported in [8] (based on strip chain be considered a limitative aspect, in certain scenarios encoders, as well) was also 0.6 mm (the chain period), but relative to item identification, tag reading at certain distance the envelope function for the sequence ‘1111…’ reported in may be necessary. In those cases, frequency-domain or that work does not provide significant dips/peaks. traditional time-domain chipless-RFID systems may be Therefore, the automatic detection of the time period (and more convenient, provided the required number of bits of hence the instantaneous velocity) by means of a (potential) the tags is not very high. However, there are applications post processing system is envisaged to be more complex. where confidence against spying or eavesdropping is Other similar encoders, based on chains of resonant necessary, e.g., product authentication and secure paper elements and used as displacement/velocity sensors have applications. The proposed system which requires tag been reported [5],[7] but the achieved space resolution reading by proximity, it is inherently robust against these (2.2 mm) is not comparable to the one reported in this work. (potentially) malicious scenarios. Thus, as in chipless-RFID Due to the small period achieved in the proposed system, the proposed approach is of interest in such encoders, the density of information per unit length is very authentication applications, in order to avoid fraud or high, particularly DPL = 16.7 bit /cm. This value is plagiarism in documents, high cost item products, etc. identical to the one reported in [8],[16], and much better than the one reported in [15], where DPL = 5 bit/cm. VI. TOLERANCE ANALYSIS However, the most relevant aspect of these encoders is the In order to study the robustness of the system against air short length of the strips. Such short length has been gap variations, and lateral or angular misalignment of the obtained thanks to the new designed reader, able to detect encoder, a tolerance analysis is needed. However, due to the the presence or absence of functional strips of such short difficulty to accurately control the involved geometrical length on top of it. The design of such reader has parameters (namely, air gap separation and lateral or represented a significant effort and constitutes the main angular displacement of the encoder), such analysis is novel aspect of this paper, as compared to previous similar carried out at simulation level. For that purpose, a 5-bit works (mainly [8],[15],[16]). The information density per sequence with alternating binary states is considered. area unit is as high as DPS = 26.04 bit/cm2, as it is inferred from the fact that the designed and fabricated 100-bit -5 Concerning the accuracy in the orthogonal direction to the -10 (a) encoder and reader planes (air gap separation), the -15 requirements are more stringent. According to the reported -20 data [Fig. 11(a)], during encoder motion, the guiding -25 system must guarantee an air gap separation comprised -30 0.17 0.19 0.21 between roughly 0.17 mm and 0.22 mm. Increasing the 0.18 reference 0.22 -35 tolerance relative to the air gap separation is possible at the -5 expense of larger encoder sizes. Nevertheless, the fabricated -10 (b) encoders have been correctly read with our in-house |S21| (dB)|f = f0 -15 experimental setup. Consequently, the reported air gap -20 tolerances seem to be reasonable for an eventual reader, -25 reference  mm  mm based on a sufficiently robust mechanical guiding system, -30  mm  mm to be used for encoder reading in a real scenario. -35 -5 VII. CONCLUSIONS -10 (c) In conclusion, high data density microwave encoders -15 based on chains of transversely oriented linear strips have -20 been proposed in this paper. The length of the strips, a -25 critical parameter determining the data density per surface reference 2 4 6 -30 1 3 5 of the encoders, has been substantially reduced, as -35 compared to previous implementations. For that purpose, a 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 dedicated reader based on a double stub loaded microstrip Displacement (mm) line has been designed and fabricated. Such reader is able to Fig. 11. Tolerance analysis. (a) Effects of the air gap variation; (b) effects detect the presence and functionality of the encoder strips of lateral misalignment; (c) effects of misalignment by encoder rotation. through near-field (capacitive) coupling, when the encoder The curves correspond to the transmission coefficient at the frequency of the interrogation signal, as the 5-bit encoder with ID code ‘10101’ is is displaced over the sensitive part of the reader (the displaced over the reader. extremes of the stubs), as needed in a reader operation. An analysis of the reader-encoder system based on the circuit The effects of the air gap variation are depicted in Fig. schematic, where the encoder strips are modeled by means 11(a). It can be appreciated that the system has been of an inter-stub coupling capacitance, has been carried out, optimized to operate with an air gap corresponding to the and the model has been validated by comparison with nominal value, where the excursion experienced by the electromagnetic simulations. Such analysis has been transmission coefficient between the two binary states is demonstrated to provide useful hints for design purposes. maximum. Slightly increasing or decreasing the air gap System validation has been carried out by reading several distance reduces such excursion. According to the fabricated encoders with different bit combinations, where the different ID codes have been achieved by cutting certain sensitivity of the transmission coefficient with the air gap strips of the encoder (tag programming). The functionality (similar to the one reported in [8]), the mechanical guiding of the system for near-field chipless-RFID applications and system for encoder motion should guarantee small for motion control (i.e., as displacement and velocity variations in the air gap separation. Further robustness sensors) has been demonstrated. Also, a tolerance analysis, against the effects of the air gap can be achieved, but at the including the effects of encoder misalignment caused by expense of larger encoder dimensions. lateral displacement and rotation, as well as the effects of The effects of lateral misalignment of the encoder are air gap variation, has been performed through shown in Fig. 11(b). According to these results, lateral electromagnetic simulation. The obtained tolerances are displacements up to 1 mm provide excursions in the reasonable in order to consider the system potentially useful transmission coefficient better than 8 dB. With this for application in a real scenario. The most relevant variation, encoder reading from the AM signal generated at achieved result is the information density per surface, as the output port of the reader in response to the interrogation high as DPS = 26.04 bit/cm2. This value, much higher than signal seems feasible. the data density obtained in other chipless-RFID systems Finally, Fig. 11(c) shows the effects of lack of alignment (including near-field, time domain, frequency domain and by encoder rotation, where the responses up to an angle of hybrid systems), has been achieved thanks to the designed 6º, with step increments of 1º, are depicted. According to and fabricated reader, able to detect very small metallic the resulting transmission coefficient variations between the elements (strips) in the encoder chain. 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Telecommunications Engineering Diploma [49] J. Machac and M. Polivka, “Influence of mutual coupling on (specializing in Electronics) and the performance of small scatterers for chipless RFID tags,” 24th Int. Telecommunications Engineering degree from the Radioelektron. Conf., 2014, pp. 1–4. Universitat Autònoma de Barcelona in 2004 and [50] M. Khaliel, A. El-Awamry, A. Fawky, M. El-Hadidy, and T. Kaiser, 2006, respectively and the PhD degree in “A novel co/cross-polarizing chipless RFID tags for high coding Electronics Engineering from the same university capacity and robust detection,” 2015 IEEE Int. Symp. Ant. Propag. & in 2012. He was Assistant Professor from 2006 to USNC/URSI National Radio Sci. Meeting, Jul. 2015, pp. 159–160. 2008 at the Universitat Autònoma de Barcelona, [51] M. Svanda, J. Machac, M. Polivka, J. Havlicek., "A comparison of where he is currently working as a Research two ways to reducing the mutual coupling of chipless RFID tag Assistant. His research interests include metamaterial concepts, passive scatterers," in Proc. of 21st International Conference on Microwave, microwaves devices, antennas and RFID. Radar and Wireless Communications (MIKON), May 2016, pp. 1-4. [52] C. Herrojo, J. Naqui, F. Paredes and F. Martín, “Spectral signature barcodes based on S-shaped Split ring resonators (S-SRR)”, EPJ Ferran Martín (M’04-SM’08-F’12) was born in Applied Metamaterials, vol. 3, pp. 1-6, Jun. 2016. Barakaldo, Spain, in 1965. He received the B.S. [53] A. Vena, E. Perret, S. Tedjini, "Chipless RFID tag using hybrid degree in physics and Ph.D. degree from the coding technique," IEEE Trans. Microw. Theory Techn., vol. 59, pp. Universitat Autònoma de Barcelona (UAB), 3356-3364, Dec. 2011. Barcelona, Spain, in 1988 and 1992, respectively. [54] A. Vena, E. Perret, S. Tedjini, “A compact chipless RFID tag using From 1994 to 2006 he was Associate Professor in polarization diversity for encoding and sensing”, 2012 IEEE Int. Electronics at the Departament d’Enginyeria Conf. RFID, pp. 191-197, 2012. Electrònica (Universitat Autònoma de Barcelona), [55] M. A. Islam and N. C. Karmakar, "A novel compact printable dual- and since 2007 he is Full Professor of Electronics. polarized chipless RFID system," IEEE Trans. Microw. Theory His research activity has been very broad, Techn., vol. 60, pp. 2142-2151, Jul. 2012. including the modelling and simulation of electron devices for high [56] I. Balbin, N.C. Karmakar, “Phase-encoded chipless RFID transponder frequency applications, millimeter wave and THz generation systems, the for large scale low cost applications”, IEEE Microw. Wireless. application of electromagnetic bandgaps to microwave and millimeter Compon. Lett., vol. 19, pp. 509-511, 2009. wave circuits, and the application of metamaterial concepts to the [57] S. Genovesi, F. Costa, A. Monorchio, G. Manara, “Chipless RFID tag miniaturization and optimization of microwave circuits and antennas. He is exploiting multifrequency delta-phase quantization encoding”, IEEE now very active in the development of microwave sensors for dielectric Ant. Wireless Propag. Lett., vol. 15, pp. 738-741, 2015. characterization and motion control, and also in the topic of chipless- [58] O. Rance, R. Siragusa, P. Lemaitre-Auger, and E. Perret, “RCS RFID. He is the head of the Microwave Engineering, Metamaterials and magnitude coding for chipless RFID based on depolarizing tag,” in Antennas Group (GEMMA Group) at UAB, and director of CIMITEC, a IEEE MTT-S Int. Microw. Symp. Dig., 2015, pp. 1–4. research Center on Metamaterials supported by TECNIO (Generalitat de [59] O. Rance, R. Siragusa, P. Lemaître-Auger, E. Perret, "Toward RCS Catalunya). He has organized several international events related to magnitude level coding for chipless RFID," IEEE Trans. Microw. metamaterials, including Workshops at the IEEE International Microwave Theory Techn., vol. 64, pp. 2315-2325, Jul. 2016. Symposium (years 2005 and 2007) and European Microwave Conference [60] C. Herrojo, J. Naqui, F. Paredes, F. Martín, “Spectral signature (years 2009, 2015, 2017, and 2018), and the Fifth International Congress barcodes implemented by multi-state multi-resonator circuits for on Advanced Electromagnetic Materials in Microwaves and Optics chipless RFID tags”, IEEE MTT-S International Microwave (Metamaterials 2011), where he has acted as chair of the Local Organizing Symposium (IMS’16), San Francisco, May 2016. Committee. He has acted as Guest Editor in several Special Issues, mainly [61] C. Herrojo, F. Paredes, J. Mata-Contreras, S. Zuffanelli and F. related to Metamaterials, in various International Journals. He has authored Martín, “Multi-state multi-resonator spectral signature barcodes and co-authored over 580 technical conference, letter, journal papers and implemented by means of S-shaped split ring resonators (S-SRR)”, book chapters, he is co-author of the book on Metamaterials entitled IEEE Trans. Microw. Theory Techn., vol. 65, no. 7, pp. 2341-2352, Metamaterials with Negative Parameters: Theory, Design and Microwave Jul. 2017. Applications (John Wiley & Sons Inc. 2008), author of the book Artificial [62] C. Feng, W. Zhang, L. Li, L. Han, X. Chen, and R. Ma, “Angle-based Transmission Lines for RF and Microwave Applications (John Wiley & chipless RFID tag with high capacity and insensitivity to Sons Inc. 2015), co-editor of the book Balanced Microwave filters (John polarization,” IEEE Trans. Ant. Propag., vol. 63, no. 4, pp. 1789– Wiley & Sons Inc. and IEEE-Press, 2018), and co-author of the book 1797, Apr. 2015. Time-Domain Signature Barcodes for Chipless-RFID and Sensing [63] A. El-Awamry, M. Khaliel, A. Fawky, M. El-Hadidy, and T. Kaiser, Applications (Springer, to be published in 2020). Ferran Martín has “Novel notch modulation algorithm for enhancing the chipless RFID generated 19 PhDs, he has filed several patents on metamaterials, and he tags coding capacity,” in IEEE Int. RFID Conf., 2015, pp. 25–31. has headed dozens of Development Contracts. [64] A. Vena, A. A. Babar, L. Sydanheimo, M. M. Tentzeris, and L. Prof. Martín is a member of the IEEE Microwave Theory and Techniques Ukkonen, “A novel near-transparent ASK-reconfigurable inkjet- Society (IEEE MTT-S). He is reviewer of the IEEE Transactions on printed chipless RFID tag,” IEEE Ant. Wireless Propag. Lett., vol. Microwave Theory and Techniques and IEEE Microwave and Wireless 12, pp. 753–756, 2013. Components Letters, among many other journals, and he serves as member [65] D.M. Pozar, Microwave Engineering, 4th Ed., John Wiley, Hoboken, of the Editorial Board of IET Microwaves, Antennas and Propagation, NJ, 2011. International Journal of RF and Microwave Computer-Aided Engineering and Sensors. He is also a member of the Technical Committees of the Cristian Herrojo was born in Barcelona, Spain, in European Microwave Conference (EuMC) and International Congress on 1983. He received the Telecommunications Advanced Electromagnetic Materials in Microwaves and Optics Technical Engineering degree in electronic systems (Metamaterials). Among his distinctions, Ferran Martín has received the and Telecommunications Engineering degree from 2006 Duran Farell Prize for Technological Research, he holds the Parc de the Universitat Autònoma de Barcelona in 2010 Recerca UAB – Santander Technology Transfer Chair, and he has been the and 2012, respectively and the PhD degree in recipient of three ICREA ACADEMIA Awards (calls 2008, 2013 and Electronics Engineering from the same university 2018). He is Fellow of the IEEE since 2012 and Fellow of the IET since in 2018. His research interests include 2016. RF/microwave devices, Chipless-RFID and RFID technology, and Metamaterials.