This is the accepted version of the journal article: Herrojo, Cristian; Paredes, Ferran; Bonache, Jordi; [et al.]. «3-D-Printed High Data-Density Electromagnetic Encoders Based on Permittivity Contrast for Motion Control and Chipless-RFID». IEEE transactions on microwave theory and techniques, Vol. 68, Issue 5 (May 2020), p. 1839-1850. DOI 10.1109/TMTT.2019.2963176 This version is available at https://ddd.uab.cat/record/258423 under the terms of the license 3D-Printed High Data-Density Electromagnetic Encoders Based on Permittivity Contrast for Motion Control and Chipless-RFID Cristian Herrojo, Member IEEE, Ferran Paredes, Member IEEE, Jordi Bonache, Member IEEE, and Ferran Martín, Fellow IEEE Abstract— 3D-printed electromagnetic encoders based on Despite the fact that it is difficult to compete against permittivity contrast are presented and discussed in this optical encoders in terms of spatial resolution (dictated by paper. The encoders, implemented by using exclusively the grid period), such encoders have two main drawbacks: dielectric materials, are based on linear chains of dielectric their cost (as well as the cost of the optical source and inclusions. Two types of encoders are considered: (i) those where the inclusions are simple apertures made on a 3D- detector), and their limited robustness when they are printed dielectric plate (or substrate), and (ii) those with subjected to certain harsh environments, e.g., with inclusions made of high dielectric constant material 3D- pollution, dirtiness, or grease, among others. The cost of the printed on a dielectric substrate (also 3D-printed). In both source/detection system can be reduced by using cases, encoding is achieved by varying the dielectric constant radiofrequencies (RF) or microwaves, rather than optical of the substrate at predefined positions by means of the signals. However, the use of the RF/microwave spectrum inclusions. For encoder reading, a microstrip line loaded with a slot resonator (etched in the ground plane) and a series gap needs a different strategy for encoder implementation. In is proposed. Such structure is very sensitive to short range [4]-[17], the proposed electromagnetic encoders were dielectric constant variations, being able to detect the presence fabricated by etching or printing a chain of metallic or absence of closely spaced inclusions when the encoder is inclusions (typically resonators or strips) on a dielectric displaced on top of the sensitive part of the reader, the slot substrate. In applications where mechanical robustness is resonator. For that purpose, the reader line is fed by a required and the cost of the encoder is not critical, e.g., harmonic signal conveniently tuned, so that an amplitude motion control, rigid substrates are preferable [4]-[7]. modulated (AM) signal containing the ID code is generated at the output port of the line. The proposed reader/encoder However, low-cost flexible substrates (including plastic or system is useful for motion control applications (linear paper) can be used in applications where the cost of the displacement and velocity sensors) and for near-field chipless- encoder is a critical issue [10]-[13]. This applies to chipless- RFID. Lower cost and major robustness against mechanical RFID, where ultra-low cost encoders (or tags) can be wearing are potential advantages of these encoders over other fabricated by printing the metallic inclusions through inkjet electromagnetic encoders based on a similar principle but printing, or by means of massive fabrication processes based on metallic inclusions. (such as rotogravure, screen-printing, offset, etc.). Although Index Terms– Microwave encoder, 3D-printing, additive the cost of commercially available conductive inks is manufacturing, chipless-RFID, motion control, microstrip moderately high, a very small quantity of ink suffices for technology, slot resonators. tag fabrication. Thus, the price of printed encoders implemented on plastic or paper substrates is situated below I. INTRODUCTION the cost of chip-based tags, and, obviously, much below the Recently, electromagnetic encoders have been proposed cost of optical encoders. as a low-cost alternative to optical encoders. In optical As compared to optical encoders, electromagnetic encoders, a grid of apertures in a metallic plate is used for encoders may offer major levels of confidence in certain coding purposes. The presence or absence of apertures industrial scenarios where the presence of contaminants or during encoder motion is detected by means of an optical pollution may jeopardize the necessary cleanness and beam, and information relative to the displacement and transparency of optical apertures. In chipless-RFID velocity (either linear or angular) of the encoder can be applications, the absence of chip in the tags (replaced by the inferred on the basis of pulse counting [1]-[3]. Optical metallic encoder), not only reduces costs, but it may also encoders with very high resolution are available in the represent a good solution in scenarios subjected to extreme market and are of high interest in many industrial temperatures or radiation (e.g., integrated circuits exhibit applications related to motion control (e.g., elevators, limited radiation hardness). However, encoders based on servomechanisms, automotive industry, etc.). printed metallic patterns may be subjected to wearing and aging effects that may limit their long-life use. Note that This paper is an expanded version from the IEEE MTT-S International wearing may be caused by unexpected contacts (due, e.g., Microwave Symposium 2019, Boston, MA, June 2-7, 2019. This work was supported by MINECO-Spain (project TEC2016-75650- to vibration) between the encoder and the sensitive part of R), by Generalitat de Catalunya (project 2017SGR-1159), by Institució the reader (a RF/microwave component able to detect the Catalana de Recerca i Estudis Avançats (who awarded Ferran Martín), and presence of metallic inclusions through near-field). by FEDER funds. C Herrojo, F. Paredes, Bonache, and F. Martín are with In [18], we proposed a new type of low-cost GEMMA/CIMITEC, Departament d’Enginyeria Electrònica, Universitat electromagnetic encoders, based exclusively on dielectric Autònoma de Barcelona, 08193 Bellaterra, Spain. E-mail: materials, and thereby robust against wearing and aging Ferran.Martin@uab.es. effects. In such encoders, the metallic chains of printed (or encoders, as compared to other encoders, is reported. etched) resonant elements or strips are replaced with Finally, the main conclusions are highlighted in Section apertures in the substrate. Since the electromagnetic VII. properties of the apertures (in particular the dielectric constant) are different from those of the substrate, such II. WORKING PRINCIPLE apertures are useful for coding purposes, as demonstrated in In the printed electromagnetic encoders based on metallic [18]. patterns reported in [4]-[17], a sensitive element able to In the work presented in [18], the functionality of all- detect the presence/absence of functional metallic elements dielectric electromagnetic encoders based on permittivity (the inclusions) in the encoder chain was used for reading contrast was demonstrated. However, the period of the purposes. Specifically, encoder reading was carried out apertures chain, similar to the size of the sensitive element through near-field, by using a transmission line based of the reader, a complementary spiral resonator (CSR), was structure, where such transmission line is typically loaded not optimized. Reduction of such period is fundamental with a resonator (or resonators) sensitive to the presence of since it dictates the sensor resolution (in the measurement metallic elements in their surroundings. of displacements and velocities) and the per unit length data Similarly, in all-dielectric encoders based on permittivity density (a key parameter in encoders based on chains of contrast, an element able to detect, through near-field, the inclusions for its use as chipless-RFID tags). In this paper, presence/absence of dielectric inclusions (on the basis of by designing (and optimizing) a completely new reader dielectric constant variation), is needed [18]. For that (i.e., the sensitive part), based on a linearly-shaped slot purpose, a high-sensitive permittivity sensor, able to detect resonator, it is demonstrated that the period of the encoders local changes in the permittivity of the encoder is a due. In can be substantially reduced. Moreover, we report not only [18], the permittivity contrast encoder was implemented by all-dielectric encoders based on apertures (with linear means of square apertures on a microwave (rigid) substrate, shape), but also permittivity contrast encoders consisting of and the sensing element of the reader was a microstrip line chains of high dielectric constant inclusions on top of a loaded with a CSR, etched in the ground plane. flexible dielectric substrate. In both cases, the encoders are In this paper, a different sensing transmission line based fully fabricated through additive manufacturing by means structure, compatible with the detection of smaller (in the of a 3D printer. All these aspects represent a significant direction of the chain axis) dielectric inclusions, is extension and added contribution with regard to the idea proposed. Nevertheless, the principle is the same; namely, and results first presented in [18]. Indeed, all the simulated the presence of the inclusions on top of the sensitive and experimental results presented in this paper, as well as element (a resonator) modifies the frequency response of the reader and the different 3D-printed encoders, are the structure. Therefore, through encoder motion, an AM original. Moreover, to the best of our knowledge, it is the signal at the output port is generated, provided the line is first time that 3D-printed electromagnetic encoders are fed with a conveniently tuned harmonic signal. The reported. envelope of such AM signal, which can be obtained by The proposed 3D-printed all-dielectric encoders are of means of an envelope detector, contains the relevant interest for motion control applications, and for near-field information, either relative to the displacement/velocity or chipless-RFID. In the former application, thick additive to the identification (ID) code. For the measurement of the manufactured encoders based on periodically located displacement and velocity, all the inclusions are present at apertures are suitable for measuring the relative position the predefined positions in the encoder chain. Thus, the and velocity between the encoder and the reader. In this quasi-instantaneous velocity is determined from the time application, system robustness (related to the rigidity and distance between adjacent peaks, or dips, in the envelope hence thickness of the encoder) is a critical aspect. By function, whereas the displacement is inferred from the contrast, in near-field chipless-RFID, encoder cost must be cumulative number of peaks, or dips. In the functionality of as much reduced as possible. For this reason, encoder the encoder as a chipless-RFID tag, only those dielectric implementation by 3D printing dielectric inclusions on a inclusions associated to the logic level ‘1’ are present. Thus narrow and flexible substrate (also 3D-printed) is the the ID code is sequentially obtained by reading the preferred option for this second application. envelope function at predefined time windows, where the The paper is organized as follows. The principle of amplitude of the envelope function determines the logic operation of the proposed all-dielectric permittivity contrast state, or bit, of the corresponding encoder position. encoders is presented in Section II. Section III focuses on The sketch of the system is depicted in Fig. 1. It is worth the design of the sensitive part of the reader, a microstrip mentioning that the frequency of the interrogation signal, fc, line loaded with slot resonator and series gap. The design is must be tuned to a value providing a large excursion in the supported by a lumped element circuit model and by full- transmission coefficient, since this enhances the modulation wave electromagnetic simulations. Section IV deals with index, hence representing major robustness against the design of the encoders. System validation is reported in misalignments between the encoder and the reader or Section V, where the functionality of the encoders as mechanical vibrations. An important difference of the displacement/velocity sensors, as well as chipless-RFID proposed system, as compared to frequency domain [19]- tags is demonstrated. This section also includes the [39], hybrid [40]-[51] and time-domain reflectometry description of the 3D-printing process and the considered (TDR) [52]-[65] chipless-RFID systems, is the fact that a materials. In section VI, a discussion related to the advantages and limitations of the proposed 3D-printed Reader Sensitive part of the reader inclusions Δx, vm , Tag ID … Z0 Envelope V0 (t) V0 P slot etched in the detector ... Vi @ fc ground plane t0 tN-1 t2 t1 t0 inclusions (apertures or ls … tN-1 εhigh or εlow materials) μstrip εhost W Encoder Pin gap li … Pout Fig. 1. Sketch of the proposed reader-encoder system. The inclusions of ws the encoder may be apertures, or high (or low) dielectric constant materials as compared with the dielectric constant of the host substrate, εhost. single harmonic signal suffices for tag reading. By contrast, frequency sweeping and a pulsed signal is required for tag reading in frequency domain (and hybrid) and TDR based Encoder wi chipless-RFID systems, respectively. This represents further Reader costs associated to the necessary electronics for the Fig. 2. Topology of the proposed reader-encoder and relevant dimensions generation of the interrogation signal and for signal (in mm). W = 3.98, ls = 24, ws = 0.2, li = 18, wi = 0.4, and P = 3.4. In the processing. Thus, in the proposed system, based on the encoder, the considered inclusions are linearly shaped apertures, but 3D- printed high (or low) dielectric constant material inclusions can be also amplitude modulation of the interrogation signal by encoder considered. motion, low cost readers are envisaged, representing a clear advantage over traditional frequency-domain and TDR- the inclusions, unable to significantly modify the based chipless-RFID systems. capacitance of the sensing resonant element (either CSR or As compared to previous electromagnetic encoders based CSRR), and consequently its resonance frequency. For this on chains of metallic elements [4]-[17], the advantage of the reason, the considered dielectric inclusions in [18] proposed all-dielectric permittivity contrast encoders lies in (apertures) exhibit a square shape and size comparable to the encoder side (rather than in the cost of the reader). Since the one of the CSR of the reader. By contrast, since the conductive inks are not used, permittivity contrast encoders capacitance of the slot resonator is mainly determined by are more robust against wearing and aging effects. the dimensions of the narrow and long slot, and by the Moreover, implementing the encoders by means of surrounding materials, it is expected that the presence of the apertures or dielectric inclusions is in general cheaper, as inclusions depicted in Fig. 2 on top of the slot resonator compared to metallic inclusions (either etched or printed significantly alters the capacitance and resonance of the through conductive inks). sensing particle (the slot). Additionally, the slot resonator exhibits higher sensitivity III. READER DESIGN AND ANALYSIS to resonance frequency variation, as compared to the one of In the use of the encoder as displacement and velocity the CSR, or CSRR. To demonstrate this, the lumped sensor, it is convenient to reduce the chain period as much element equivalent circuit models of the slot- and CSR- as possible, since such period dictates the spatial resolution. loaded (or CSRR-loaded) microstrip line are considered Thus, a sensitive element able to detect dielectric constant (Fig. 3). In these models, Lc and Cc are the inductance and variations within small ranges is required. Note that a small capacitance, respectively, of the bare resonators (i.e., encoder period is also convenient for identification without any material on top of it, or surrounding it), and C purposes, in order to enhance the data density per unit describes the coupling capacitance between the line and the length (DPL). In this work, the proposed sensing element is CSR, or CSRR, in the corresponding circuit model. If we a microstrip line loaded with a transverse (linearly shaped) now consider that the resonant element is covered by a slot resonator and with a series capacitive gap. The slot material under test (MUT), with dielectric constant MUT, resonator is very sensitive to variations in the dielectric and that the thickness of such material is significantly larger constant of the material in contact with it, or surrounding it. than the transverse dimensions of the slots, the resonance Moreover, its elongated shape is very interesting in order to frequency is given by [66] consider dielectric encoder inclusions of similar shape, 1 𝑟 +𝑀𝑈𝑇 −1/2 thereby providing short dimensions in the direction of the 𝑓𝑜 = {𝐿𝑐 (𝐶 + 𝐶𝑐 )} (1) 2 𝑟 +1 chain axis, as required in near-field chipless-RFID systems with sequential bit reading. The topologies of the reader and where r is the dielectric constant of the substrate (note that encoder with all inclusions present at their predefined C = 0 pF for the slot-loaded line). The validity of (1) is also positions are depicted in Fig. 2. restricted to a substrate thickness much larger than Note that the type of encoders depicted in Fig. 2 cannot transverse slot dimensions. These limitations relative to a be easily read with a sensing element based on a microstrip minimum substrate and MUT thickness are necessary to line loaded with a CSR, as it was carried out in [18], or with guarantee that the electric field lines generated in the slots a complementary split ring resonator (CSRR). The reason is do not reach the opposite interface of the substrate or MUT, the elongated shape of The solution to this problem is to add an element to the (a) Zo, kl Cc Zo, kl slot-loaded line, i.e., a series gap, as illustrated in Fig. 2. Pin Pout The circuit model including the series gap is depicted in Fig. 4, where Cs is the capacitance of the gap and Cf Lc accounts for the fringing capacitances. By including the (b) series gap, a reflection zero (or transmission peak) located to the left of the resonance frequency of the slot is Zo, kl L L Zo, kl Pin Pout introduced. The reflection zero frequency is the one C providing an image impedance, ZI, identical to the reference impedance of the ports, Z0. For a -network with series impedance Zs and shunt impedance Zp, the image Lc Cc impedance is given by [67] 𝑍𝑠 𝑍𝐼 = 𝑍𝑝 √ (5) 𝑍𝑠 + 2𝑍𝑝 Fig. 3. Circuit model of the slot-loaded line (a) and CSR- or CSRR-loaded and the condition for total transmission (ZI = Z0) is line (b). Z0 and kl are the characteristic impedance and electrical length, respectively, of the access lines. 𝑍02 𝑍𝑝 𝑍𝑠 = 2 (6) a necessary condition to express the slot capacitance of the 𝑍𝑝2 − 𝑍02 resonator as shown in (1). In general, the presence of a thick In general, the fringing capacitance is small, providing a enough MUT (or dielectric inclusion, the case of actual large value of Zp. Thus, the reflection zero (or transmission interest in this work) cannot be ensured. Nevertheless, this peak) frequency can be approximated by the frequency does not represent a loss of generality concerning the satisfying Zs = 0, that is conclusions of the present analysis. 1 𝑟 + 𝑀𝑈𝑇 −1/2 The relative sensitivity of resonance frequency with 𝑓𝑟 = {𝐿𝑐 (𝐶𝑠 + 𝐶𝑐 )} (7) variations in the dielectric constant of the MUT is given by 2 𝑟 + 1 1 𝑑𝑓0 By choosing a small value of the capacitance Cs, the 𝑆= (2) transmission peak can be situated very close to the 𝑓0 𝑑 𝑀𝑈𝑇 transmission zero. According to (1) and (7), the presence of and after some simple algebra, the following result is a MUT in contact to the slot resonator decreases both obtained: frequencies. Consequently, by setting the interrogation signal frequency to the peak frequency of the line loaded 1 𝐶𝑐 with the bare slot (i.e, fc = frMUT = 1), it is expected to achieve 𝑆=− (3) 2 𝐶𝑐 (𝑟 + 𝑀𝑈𝑇 ) + 𝐶(𝑟 + 1) a large excursion of the transmission coefficient when the slot resonator is loaded with a certain MUT (even in the According to (3), the sensitivity increases by reducing C, case that the MUT is separated from the slot by an air gap, and therefore it can be concluded that it is optimized in the as required in a real scenario to avoid mechanical friction). slot-loaded line (where C = 0, as mentioned). Indeed, in this By this means, the Q-factor of the slot resonator is not case, the relative sensitivity is found to be determined only altered, but an abrupt transition in the response to the left of by the dielectric constants of the substrate and MUT, the transmission zero is achieved. according to 1 In the designed reader, the reflection and transmission 𝑆=− (4) zero frequencies with unloaded slot resonator are set to 2(𝑟 + 𝑀𝑈𝑇 ) fr = 3.85 GHz and f0 = 4.49 GHz, respectively. This does and it increases by decreasing the dielectric constant of the not determine univocally the element values of the circuit substrate (MUT is not a design parameter). model of the slot-loaded line with series gap, but this is an For our purposes, we do not only need high frequency advantage, rather than a drawback, since extreme sensitivity, but also a high Q-factor. The reason is that the geometrical values can be avoided. For the design of the excursion of the transmission coefficient at the design sensitive part of the reader, the following procedure has (interrogation) frequency, which depends on both been followed: we have set the width of the slot to the parameters, is expected to be enhanced by increasing S and minimum value (a convenient strategy to minimize the Q. Note that a strong variation of the resonance frequency period of the dielectric inclusions, or encoder, chain). To with MUT (corresponding to high values of S) is not enhance the Q-factor, the inductance of the slot should be efficient in terms of transmission coefficient variation at a minimized since the bandwidth of the slot resonator is fixed frequency, if the bandwidth of the resonance is high proportional to Lc/Cc. For this reason, the considered (low Q-factor). Similarly, a high Q-factor (narrow notches) topology of the sensitive resonator is a linear slot, rather does not suffice to guarantee a minimum excursion of the than a dumbbell-shaped defect ground structure (DB-DGS). transmission coefficient, if S is small. Thus, both S and Q By tuning the slot length, we are able to vary its resonance need to be as large as possible. Decreasing r enhances S, frequency. Therefore, the length of the slot is dictated by but it also reduces the Q-factor, as far as the resonator the required resonance frequency, f0 = 4.49 GHz, as capacitance decreases. indicated before. Then the gap distance is tuned until (i) the 0 fc (a) Cc Pin Pout wi=2.6 mm |S21| (dB) -10 Cs Lc wi=0 mm Cf Cf -20 -30 Fig. 4. Circuit model of the slot-loaded line with series gap. Access lines 3.0 3.5 4.0 4.5 5.0 are excluded in this model. Frequency (GHz) (a) 0 -5 (b) -10 S11 |S21| (dB)|f = fc -10 -20 S21 -15 |S11|, |S21| (dB) -30 EM sim. bare reader aperture wi -40 Meas. bare reader -20 slot 0 Dielectric slab -25 -10 0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 S11 -20 wi (mm) S21 -30 EM sim. loaded reader Fig. 6. Simulated frequency responses of the reader with a single aperture, -40 Meas. loaded reader obtained by varying the aperture width (a), and variation of the transmission coefficient at the operating frequency (b). The considered air 3.2 3.6 4.0 4.4 4.8 gap is 0.2 mm. Frequency (GHz) (b) 0 reader substrate is the Rogers RO4003 with thickness h = 1.524 mm, dielectric constant r = 3.55 and loss tangent |S11|, |S21| (dB) -15 tan = 0.0021). Figure 5(b) depicts the response of the S11 unloaded structure inferred from lossless full-wave S21 -30 electromagnetic simulation using Keysight Momentum. This EM sim. response is compared with the response obtained by circuit -45 Circ. sim. simulation of the lumped model (also without losses), where the extracted parameters (indicated in the caption) 100 have been considered. The agreement is very good, pointing S S () 50 out the validity of the model. Then we have simulated  (including losses) the response by considering the slot 0 loaded with a dielectric slab of thickness 0.635 mm, -50 dielectric constant MUT = 10.2 and loss tangent  -100 tan = 0.0027 (corresponding to the commercial Rogers 3.2 3.6 4.0 4.4 4.8 RO3010 substrate with such parameters), separated 0.2 mm (air gap) from the slot [Fig. 5(a)]. As expected, the overall Frequency (GHz) response is shifted downwards, and a significant excursion of the transmission coefficient at the operating frequency, Fig. 5. Simulated (lossy) and measured responses of the bare reader and indicated in the caption of Fig. 5, is achieved. reader loaded with a dielectric slab with the dielectric constant and thickness indicated in the text (a), and frequency response of the bare reader inferred from lossless electromagnetic and circuit simulation of Fig IV. ENCODER DESIGN 4 (b). Extracted element parameters are Cs = 0.17 pF, Lc = 1.24 nH, In a first step towards the experimental validation of the Cc = 1.11 pF, Cf = 47.6 fF. In (a), the excursion experienced by the insertion loss by loading the slot at the operating frequency fc = frMUT = 1 is proposed reader/encoder system (to be discussed in the next 14.3 dB. section), we have considered an all-dielectric encoder implemented by opening narrow apertures on the MUT of reflection zero with bare reader, frMUT = 1, is located as close the previous section (Rogers RO3010 substrate, with as possible to the transmission zero, and (ii) the value of the h = 0.635 mm and r = 10.2). In order to determine the transmission coefficient at fc = frMUT = 1 is around −5 dB dimensions of the apertures and their separation, we have [Fig. 5(a)]. Such value is reasonable in order to obtain a carried out a set of electromagnetic simulations including large excursion of the transmission coefficient experienced losses. In the first simulations, we have considered a single by tag motion. It has been found that by setting aperture of 24 mm length (identical to the length of the slot frMUT = 1 = 3.85 GHz, the value of −5 dB for the insertion resonator), and we have simulated the frequency responses loss is roughly obtained. that result by varying the aperture width, with perfectly With the previous procedure, the dimensions indicated in aligned aperture and slot. The results are depicted in Fig. the caption of Fig. 2 have been obtained (the considered 6(a), whereas Fig. 6(b) shows the variation of the (a) 0 fc -5 '1' '1' '1' '1' -10 |S21| (dB)|f = fc |S21| (dB) -10 li=24 mm li=18 mm -15 -20 -20 -25 4-aperture encoder implemented on Rogers substrate -30 4-MT-copper inclusions on PLA substrate 3.0 3.5 4.0 4.5 5.0 -30 Frequency (GHz) 0 2 4 6 8 10 12 -6 Displacement (mm) (b) -8 wi = 0.4 mm Fig. 9. Variation of the transmission coefficient at the operating frequency |S21| (dB)|f = fc -10 obtained by displacing a 4-aperture encoder implemented on commercial -12 microwave substrate over the reader at 0.2 mm distance (air gap), and by -14 displacing a 4-inclusion (of MT-copper) on PLA substrate over the reader aperture wi=0.4 mm -16 slot at 0.075 mm distance. -18 Dielectric slab li length from the nominal value (in all the cases the width set -20 to 0.4 mm). The results, depicted in Fig. 7, indicate that 18 19 20 21 22 23 24 length reduction has negligible effect on the response, at li (mm) least within the considered range. Therefore, the length of the apertures has been set to 18 mm, i.e., 6 mm shorter than Fig. 7. Simulated frequency responses of the reader with a single aperture, the slot length. obtained by varying the aperture length (a), and variation of the transmission coefficient at the operating frequency (b). The considered air Finally, in the third subset of electromagnetic gap is 0.2 mm. simulations, we have analyzed the effects of aperture separation by considering a pair of apertures with the 0 fc previously indicated dimensions, and variable distance. In (a) this case, the two apertures are situated equidistantly from the slot resonator. The responses obtained from |S21| (dB) -10 s=0.2 mm s=4.2 mm electromagnetic simulation by increasing the aperture -20 separation are shown in Fig. 8(a), whereas Fig. 8(b) depicts the transmission coefficient at fc. As expected, by increasing the aperture separation, the response tends to the one of the -30 reader loaded with a uniform substrate. Reducing the 3.0 3.5 4.0 4.5 5.0 apertures separation favors encoder resolution and data Frequency (GHz) density, but at the expense of smaller dynamic range, (or (b) wi=0.4 mm excursion of the transmission coefficient at fc). Thus, we 0 aperture1 s have set the aperture separation to 3 mm. With this value, |S21| (dB)|f = fc aperture2 the period of the apertures chain is 3.4 mm, and the -10 transmission coefficient at fc is around −21 dB [see li = 18 mm Dielectric slab wi = 0.4 mm, li = 18 mm Fig. 8(b)]. This provides a variation in the transmission -20 coefficient at the operating frequency (with and without -30 aperture on top of the slot resonator) of roughly 14 dB, as corroborated in Fig. 9. This figure depicts the simulated 0.3 0.6 0.9 1.2 1.5 1.8 2.1 2.4 2.7 3.0 3.3 3.6 3.9 4.2 transmission coefficient at fc, obtained by displacing a 4- s (mm) aperture encoder over the slot resonator. This excursion of the transmission coefficient is enough to provide a high Fig. 8. Simulated frequency responses of the reader with a pair of modulation index, as required to read the encoder. apertures, obtained by varying the aperture separation (a), and variation of the transmission coefficient at the operating frequency (b). The considered air gap is 0.2 mm. V. EXPERIMENTAL VALIDATION transmission coefficient at the operating frequency, fc. As A. Validation with Aperture Based Encoder Implemented the aperture widens, the response shifts up, towards the on Commercial Microwave Substrate response corresponding to the bare reader. It is necessary to The photograph of the fabricated reader and the one of a choose a value of the aperture width as small as possible, fabricated 50-aperture encoder (implemented on the Rogers but providing a value of the transmission coefficient at fc as RO3010 substrate with thickness 0.635 mm, dielectric close as possible to the value of the bare reader. Taking this constant MUT = 10.2 and loss tangent tan = 0.0027) are tradeoff into account, and in view of Fig. 6(b), we have set depicted in Fig. 10. In this first prototype encoder, all the the aperture width to 0.4 mm. apertures are present, since its potential application (taking In order to determine the length of the apertures, we have into account the non-negligible cost of the considered carried out electromagnetic simulations by decreasing the encoder material) is the measurement of the relative (a) Reader (bottom view) gap slot RS Pro Mt-Copper apertures Fig. 13. Photograph of the 3D-printed encoder based on 15 apertures. The Reader (top view) thickness of the printed slab is 0.65 mm. (b) 50-aperture encoder 0.5 0.4 V0 (V) Fig. 10. Photograph of the fabricated reader (a) and encoder (b). 0.3 RFout Isolator 0.2 Tm Vout Schottky vm = 14P/Tm= 20.02 mm/s 0.1 diode Signal 3.5 4.0 4.5 5.0 5.5 6.0 generator Time (s) Fig. 14. Measured envelope function for the 15-aperture 3D-printed encoder. (model Agilent MSO-X 3104A) is used to visualize the Stepper motor Reader envelope function of the encoders. The relative motion between the reader and the encoder is achieved by means of RFin a linear stepper motor (model THORLABS LTS300/M). Lastly, the envelope detector has been implemented by means of a Schottky diode (model Avago HSMS-2860), an Fig. 11. Experimental setup used for encoder reading. active prove (model Agilent N2795A) with resistance and capacitance R = 1 MΩ and C = 1 pF, respectively, and an 1.0 0.4 vm = 10P/Tm= 20.09 mm/s isolator (model ATM PNR ATc4-8) in order to prevent from V0 (V) 0.8 0.2 unwanted reflections from de diode. The measured 0.0 Tm envelope function is depicted in Fig. 12. From the distance 0.6 between adjacent peaks, or dips (both perfectly visible), the V0 (V) 4.2 4.5 4.8 5.1 5.4 5.7 6.0 0.4 Time (s) relative velocity between the encoder and the reader is 0.2 inferred, provided the encoder period is well known. The displacement from a reference position is given by the 0.0 cumulative number of periods in the envelope function, 4 5 6 7 8 9 10 11 12 measured from that position. According to the results of Time (s) Fig. 12, the averaged velocity over 10 periods has been found to be 20.09 mm/s, i.e., very similar to the nominal Fig. 12. Measured envelope function for the fabricated 50-aperture value (20 mm/s). encoder. B. 3D-Printing Process and Materials displacement and velocity between the reader and the Let us now consider the implementation of 3D-printed encoder. encoders. For encoder fabrication, the Ultimaker 3 Before encoder reading, we have measured the response Extended 3D printer has been used. Fused filament of the fabricated bare reader and the one with the fabrication (FFF) is the manufacturing technology used in considered substrate on top of the slot resonator (at 0.2 mm such 3D printer. This is a 3D-printing process that uses a from it). Such responses, included in Fig. 5, show good continuous filament of a thermoplastic material. According agreement with the lossy electromagnetic simulations. to specifications [68], this 3D-printer provides a lateral and Particularly, the simulated and measured reflection zero vertical resolution of 12.5 m and 2.5 m, respectively. frequencies coincide to a good approximation. PLA Polylactic acid and RS Pro MT-Copper are the For encoder reading, the frequency of the interrogation considered filaments used to fabricate the encoders. Before signal has been set to such (measured) reflection zero encoder fabrication, the electromagnetic properties of such frequency, i.e., fc = 3.85 GHz. As shown in Fig. 11, such filaments, i.e. the dielectric constant and the loss tangent, interrogation signal is generated by means of a function have been determined. To this end, the resonant cavity generator (model Agilent E4438C) while an oscilloscope Agilent 85072A has been used. Square samples of 60 mm -5 0.075 mm 0.100 mm 0.125 mm 0.150 mm 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 0.7 |S21| (dB)|f = fc -10 0.6 0.5 -15 1 1 0 0 0 1 1 1 0 0 1 1 1 1 1 -20 0.7 0 1 2 3 4 5 6 V0 (V) 0.6 Displacement (mm) 0.5 Fig. 15. Effects of the air gap variation on the transmission coefficient at fc for the encoder based on MT-copper inclusions on PLA substrate. For time-consuming reasons, this tolerance analysis has been carried out 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 through simulation by considering 2-MT copper inclusions on PLA 0.7 substrate. 0.6 Code 110001110011111 0.5 RS Pro Mt-Copper 5.1 5.4 5.7 6.0 6.3 6.6 6.9 7.2 7.5 inclusions (εr = 7.6) Time (s) PLA Polylactic acid (εr = 3) Fig. 17. Measured envelope functions for the 3D-printed encoders based on high dielectric constant inclusions (MT-copper). Code 111111111111111 Code 101010101010101 C. Validation with Aperture Based 3D-Printed Encoders Similar to the encoder shown in Fig. 10, in the first 3D- printed encoder implementation, the inclusions are also apertures, but made on a 3D-printed dielectric plate during fabrication. In this case, the considered filament is RS Pro Fig. 16. Photograph of 15-bit encoders with the indicated ID codes. The dimensions of the inclusions are length 18 mm and width 0.4 mm, whereas MT-Copper. Since the value of the dielectric constant of their separation is 3 mm (corresponding to an encoder period of 3.4 mm). this material is not very different from the one of the The estimated thickness of the 3D-printed substrate is 0.2 mm, and the dielectric substrate of the encoder of Fig. 10, we expect that estimated thickness of the dielectric inclusions is 0.2 mm. by fabricating 3D-printed encoders with identical aperture dimensions and separation (Fig. 13), the same reader can be side length with a thickness of 1 mm have been 3D-printed useful for reading purposes. Figure 14 depicts the envelope with both types of filaments (these slab dimensions are function of the 3D-printed 15-aperture encoder that has those recommended by Agilent). The measured dielectric been inferred by using the reader of Fig. 10, as well as the constant and loss tangent of PLA Polylactic acid are εr = 3 experimental setup used for reading the previous encoder. and tanδ = 0.010, respectively, whereas for RS Pro MT- The presence of the apertures is clearly discerned as peaks Copper the values that have been obtained are εr = 7.6 and in the envelope function. Therefore, the functionality of tanδ = 0.015. As will be shown next, such dielectric these 3D-printed encoders is demonstrated. It is important constant values are adequate to implement the two types of to highlight that despite the fact that the encoder materials 3D-printed permittivity contrast encoders considered in this are different than those of the encoder of Fig. 10, it has not paper. We would like to mention that the resonant cavity been necessary to redesign the sensitive part of the reader. Agilent 85072A is especially suited for the measurement of This is an important aspect related to system versatility, the permittivity and loss tangent of low-loss and thin film understood as the capability to use the reader in a variety of materials. Thus, for RS Pro MT-Copper (with copper scenarios, including different types of encoders. content of 80% according to specifications), the loss tangent value inferred from the resonant cavity may be D. Validation with 3D-Printed Encoders Based on questionable. For this reason, we have compared the Dielectric Inclusions estimated loss tangent obtained by the resonant cavity with In the second 3D-printed encoder realization, the the one determined by means of a non-resonant inclusions are made of a high dielectric constant material transmission-line based method [69]. The measured loss 3D-printed on a dielectric substrate (also 3D-printed). Thus, tangent of RS Pro MT-Copper provided in [69] is roughly the substrate has been printed by using PLA Polylactic acid, 0.016, which is in good agreement with the value obtained whereas the inclusions are strips of printed RS Pro MT- by means of the resonant cavity. Nevertheless, regardless of Copper. Before encoder implementation, we have simulated the loss factor and dielectric constant of RS Pro MT- the transmission coefficient at fc, achieved by displacing a Copper, the functionality of the fabricated encoders is 4-inclusions encoder over the slot resonator at 0.07 mm demonstrated, as it will be shown in the next subsections. distance (air gap). The result, depicted in Fig. 9, indicates that by reducing the airgap to 0.075 mm, it is possible to This period is not as small as the one achieved in encoders obtain a significant excursion of the transmission based on chains of transversally oriented metallic strips coefficient, necessary to properly read the encoder. In fact, [15],[16], where a period as small as 0.6 mm was achieved. the most important parameter influencing the performance The reason is that metals are more efficient than dielectrics of the proposed system is the air gap distance between the in modifying the capacitance of a resonant element through reader and the encoder. As the air gap increases, obviously, non-contact. Namely, since dielectrics do not “absorb” the the reader is more insensitive to the presence of the electric field lines, as metals do, they need more spatial encoder, and the dynamic range decreases. The effects of extension than metals in order to produce the same the air gap variation on the excursion of the transmission capacitance (and hence resonance frequency) variation. coefficient are depicted in Fig. 15. According to these According to the previous words, the reported all- results, the guiding system for encoder motion should dielectric permittivity contrast encoders cannot compete ensure a maximum air gap of the order of 0.1 mm while the against encoders based on metallic inclusions (etched or encoder is being displaced over the sensitive part of the printed) in terms of resolution and data density. However, reader (this seems to be feasible in a real scenario). metallic elements are sensitive to wearing and aging effects, The photograph of a set of 3D-printed 15-bit encoders potentially caused by chafing, rubbing, etc. Thus, the with different ID codes is depicted in Fig. 16, where reported encoders are a good option in applications where a dimensions are indicated (see caption). We have read these long-life use is required, at the expense of more limited encoders by means of the designed reader and experimental performance as compared to metallic-based encoders. setup available in our laboratory (indicated before). The Concerning encoder cost, the estimated material cost of results, depicted in Fig. 17, reveal that the different the 3D-printed encoder based on apertures is 0.05 € (using encoders are correctly read. Note, however, that in these MT-copper), whereas the material cost of the one fabricated encoders the permittivity contrast is complementary to the on the Rogers substrate is 4.78 €. On the other hand, with one of the previous encoders; that is, the inclusions are regard to the encoders based on dielectric inclusions, the made of a high dielectric constant material, whereas the cost of the materials (PLA Polylactic acid and RS Pro MT- dielectric constant of the substrate is comparatively small. Copper) is roughly two orders of magnitude cheaper than By contrast, in the previous encoders, the inclusions are the cost of conductive inks (in this case the comparison apertures (with r =1) made on a high dielectric constant should be made with tags based on printed metallic material (substrate). For this reason, the presence of inclusions), since the cost of 1 kg of ceramic based inclusions on top of the slot resonator of the reader provides filaments used and the DupontTM PE410 conductive ink a dip, rather than a peak, in the envelope functions (see Fig. (which was used in [13] by the authors) are around 80 € and 17), contrary to the encoders based on apertures (see Figs. 14000 €, respectively. However, inkjet-printed metallic tags 12 and 14). For this same reason, the response in Fig. 9 on ordinary paper substrates (as demonstrated in [13]) are corresponding to the encoder based on MT-copper very cheap, owing to the cost of ordinary paper, and to the inclusions is complementary to the one of the encoder based fact that an extremely small amount of ink, i.e., a single on apertures. layer, suffices for tag functionality. Nevertheless, the estimated material cost of the 3D-printed 15-bit encoder VI. DISCUSSION based on dielectric inclusions is 0.04 €, while the estimated As compared to the all-dielectric encoders reported in material cost of 15-bit inkjet-printed encoder on ordinary [18], the proposed encoders exhibit a smaller period. This paper is 0.12 €. This aspect demonstrates that all-dielectric has direct impact on spatial resolution, in the application of encoders are cheaper than encoders based on metallic the reader/encoder system as linear displacement sensor. inclusions. Specifically, the cost of all-dielectric encoders Concerning the data density per unit surface (DPS) and per corresponds to a reduction of roughly 67% of the cost of unit length (DPL), the achieved values are DPS = 1.66 metallic-inclusion based encoders. bit/cm2 and DPL = 2.99 bit/cm (50% higher as compared to The implementation of 3D-printed all-dielectric encoders the design proposed in [18]), respectively. As discussed in on paper substrate is left for future works. Nevertheless, it previous papers [15],[16], the DPL (rather than the DPS) is should be taken into account that 3D-printing encoding can the key data density parameter in encoders based on chains be directly carried out during manufacturing of 3D-printed of inclusions, and intended to be used as near-field chipless- products, e.g., by generating an ID code based on apertures RFID tags. The reason is that the shape factor is extremely or on dielectric inclusions. This does not represent an extra elongated if the number of bits is high and the DPL is production cost, and, at the same time, the use of a moderate or small. Therefore, the reduction of the period of dedicated tag is avoided. Obviously, the use of optical all-dielectric encoders is essential for both applications. For barcodes is a very low cost alternative (extremely extended that purpose, the strategy has been to consider a reader able nowadays). However, such barcodes may be subjected to to detect the presence or absence of very small inclusions in wearing, can be manipulated, and their data capacity is the direction of the chain axis (i.e., transversally oriented limited. By contrast, in the proposed encoders, the number linear apertures or dielectric strips). Note that the width of of bits is only limited by encoder size and encoder the inclusions in the encoders reported in this paper is as implementation in the considered item or product offers small as 0.4 mm. Nevertheless, the required separation very high robustness against mechanical wearing and between the inclusions, for proper discernment of their exposure to harsh environments (e.g., contacting the presence or absence, is 3 mm (as it has been justified in the products with water or liquids does not deteriorate the previous section), providing an encoder period of 3.4 mm. encoder). Thus, the reported 3D-printing approach for encoder fabrication may be of interest in various scenarios. rotors loaded with multiple chains of split ring resonators (SRRs)”, IEEE Sensors J., vol.18, no. 17, pp. 7055-7065, Sep. 2018. To the best of our knowledge, it is the first time that 3D- [8] C. Herrojo, J. Mata-Contreras, F. Paredes, Ferran Martín, “Near-Field printed all-dielectric electromagnetic encoders based on Chipless RFID Encoders with Sequential Bit Reading and High Data permittivity contrast are reported. Capacity”, IEEE MTT-S Int. Microw. Symp. (IMS’17), Honolulu, Hawaii, Jun. 2017. [9] C. Herrojo, J. Mata-Contreras, F. Paredes, F. Martín, “High data VII. CONCLUSIONS density and capacity in chipless radiofrequency identification In conclusion, full 3D-printed all-dielectric (chipless-RFID) tags based on double-chains of S-shaped split ring resonators (S-SRRs)”, EPJ Appl. Metamat., vol. 4, article 8, 6 pages, electromagnetic encoders based on permittivity contrast Oct. 2017. have been reported in this paper. The encoders consist of a [10] C. Herrojo, J. Mata-Contreras, F. Paredes, Ferran Martín, “Near-field chain of dielectric inclusions on a host dielectric material chipless RFID system with high data capacity for security and with significantly different dielectric constant. We have authentication applications”, IEEE Trans. Microw. Theory Techn., vol. 65 (12), pp. 5298-5308, Dec. 2017. considered two types of encoders: (i) those where the [11] C. Herrojo, J. Mata-Contreras, F. Paredes, A. Núñez, E. Ramon, and inclusions are simple apertures, and (ii) those where the F. 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Sensors J., to be published, doi: 10.1109/JSEN.2019.2941050. [47] C. Herrojo, J. Naqui, F. Paredes, F. Martín, “Spectral signature barcodes implemented by multi-state multi-resonator circuits for Cristian Herrojo was born in Barcelona, Spain, in chipless RFID tags”, IEEE MTT-S International Microwave 1983. He received the Telecommunications Symposium (IMS’16), San Francisco, May 2016. Technical Engineering degree in electronic systems [48] C. Herrojo, F. Paredes, J. Mata-Contreras, S. Zuffanelli and F. and Telecommunications Engineering degree from Martín, “Multi-state multi-resonator spectral signature barcodes the Universitat Autònoma de Barcelona in 2010 implemented by means of S-shaped split ring resonators (S-SRR)”, and 2012, respectively and the PhD degree in IEEE Trans. Microw. Theory Techn., vol. 65, no. 7, pp. 2341-2352, Electronics Engineering from the same university Jul. 2017. in 2018. His research interests include [49] C. Feng, W. Zhang, L. Li, L. Han, X. Chen, and R. Ma, “Angle-based RF/microwave devices, Chipless-RFID and RFID chipless RFID tag with high capacity and insensitivity to technology, and Metamaterials. polarization,” IEEE Trans. Ant. Propag., vol. 63, no. 4, pp. 1789– 1797, Apr. 2015. Ferran Paredes was born in Badalona [50] A. El-Awamry, M. Khaliel, A. Fawky, M. El-Hadidy, and T. Kaiser, (Barcelona), Spain in 1983. He received the “Novel notch modulation algorithm for enhancing the chipless RFID Telecommunications Engineering Diploma tags coding capacity,” in IEEE Int. RFID Conf., 2015, pp. 25–31. (specializing in Electronics) and the [51] A. Vena, A. A. Babar, L. Sydanheimo, M. M. Tentzeris, and L. Telecommunications Engineering degree from the Ukkonen, “A novel near-transparent ASK-reconfigurable inkjet- Universitat Autònoma de Barcelona in 2004 and printed chipless RFID tag,” IEEE Ant. Wireless Propag. Lett., vol. 2006, respectively and the PhD degree in 12, pp. 753–756, 2013. Electronics Engineering from the same university [52] A. Chamarti and K. Varahramyan, "Transmission delay line based ID in 2012. He was Assistant Professor from 2006 to generation circuit for RFID applications," IEEE Microw. Wireless 2008 at the Universitat Autònoma de Barcelona, Compon. Lett., vol. 16, pp. 588-590, 2006. where he is currently working as a Research [53] J. Vemagiri, A. Chamarti, M. Agarwal, and K. Varahramyan, Assistant. His research interests include metamaterial concepts, passive “Transmission line delay-based radio frequency identification (RFID) microwaves devices, antennas and RFID. Jordi Bonache was born in Cardona (Barcelona), Spain, in 1976. He received the Physics and Electronics Engineering degrees and the Ph.D. in electronics engineering from the Universitat Autònoma de Barcelona, in 1999, 2001 and 2007, respectively. In 2000, he joined the High Energy Physics Institute of Barcelona, where he was involved in the design and implementation of the control and monitoring system of the MAGIC telescope. In 2001, he joined the Department of Electronics Engineering, Universitat Autònoma de Barcelona, where he is currently Professor of Electronics. From 2006 to 2009, he was an Executive Manager with CIMITEC, Bellaterra, Spain, where he is currently a Project Manager and leads the research on RFID and antennas. His current research interests include active and passive microwave devices, metamaterials, antennas and RFID. Ferran Martín (M’04-SM’08-F’12) was born in Barakaldo, Spain, in 1965. He received the B.S. degree in physics and Ph.D. degree from the Universitat Autònoma de Barcelona (UAB), Barcelona, Spain, in 1988 and 1992, respectively. From 1994 to 2006 he was Associate Professor in Electronics at the Departament d’Enginyeria Electrònica (Universitat Autònoma de Barcelona), and since 2007 he is Full Professor of Electronics. His research activity has been very broad, including the modelling and simulation of electron devices for high frequency applications, millimeter wave and THz generation systems, the application of electromagnetic bandgaps to microwave and millimeter wave circuits, and the application of metamaterial concepts to the miniaturization and optimization of microwave circuits and antennas. He is now very active in the development of microwave sensors for dielectric characterization and motion control, and also in the topic of chipless- RFID. He is the head of the Microwave Engineering, Metamaterials and Antennas Group (GEMMA Group) at UAB, and director of CIMITEC, a research Center on Metamaterials supported by TECNIO (Generalitat de Catalunya). He has organized several international events related to metamaterials, including Workshops at the IEEE International Microwave Symposium (years 2005 and 2007) and European Microwave Conference (years 2009, 2015, 2017, and 2018), and the Fifth International Congress on Advanced Electromagnetic Materials in Microwaves and Optics (Metamaterials 2011), where he has acted as chair of the Local Organizing Committee. He has acted as Guest Editor in several Special Issues, mainly related to Metamaterials, in various International Journals. He has authored and co-authored over 580 technical conference, letter, journal papers and book chapters, he is co-author of the book on Metamaterials entitled Metamaterials with Negative Parameters: Theory, Design and Microwave Applications (John Wiley & Sons Inc. 2008), author of the book Artificial Transmission Lines for RF and Microwave Applications (John Wiley & Sons Inc. 2015), co-editor of the book Balanced Microwave filters (John Wiley & Sons Inc. and IEEE-Press, 2018), and co-author of the book Time-Domain Signature Barcodes for Chipless-RFID and Sensing Applications (Springer, to be published in 2020). Ferran Martín has generated 19 PhDs, he has filed several patents on metamaterials, and he has headed dozens of Development Contracts. Prof. Martín is a member of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S). He is reviewer of the IEEE Transactions on Microwave Theory and Techniques and IEEE Microwave and Wireless Components Letters, among many other journals, and he serves as member of the Editorial Board of IET Microwaves, Antennas and Propagation, International Journal of RF and Microwave Computer-Aided Engineering and Sensors. He is also a member of the Technical Committees of the European Microwave Conference (EuMC) and International Congress on Advanced Electromagnetic Materials in Microwaves and Optics (Metamaterials). Among his distinctions, Ferran Martín has received the 2006 Duran Farell Prize for Technological Research, he holds the Parc de Recerca UAB – Santander Technology Transfer Chair, and he has been the recipient of three ICREA ACADEMIA Awards (calls 2008, 2013 and 2018). He is Fellow of the IEEE since 2012 and Fellow of the IET since 2016.